Adapt the discussion at the end of Sec. 57 to show that the number of rectangular-waveguide modes whose cutoff frequencies are less than a given frequency \(\omega_{\max } x\) approximately, $$ N=\frac{\omega_{\max }^{2}}{2 \pi c^{2}} a b, $$ where \(N\) is assumed to be very large, and hence that the density of modes per unit frequency interval \(d N / d \omega\) increases linearly with frequency. Hint: Count both TE and TM modes.

Short Answer

Expert verified
Answer: The density of modes per unit frequency interval in a rectangular waveguide increases linearly with frequency.

Step by step solution

01

Write down the formulas for the cutoff frequencies

The cutoff frequency \(\omega_c\) in a rectangular waveguide of dimensions \(a\) and \(b\) is given by the following equations for TE and TM modes. \begin{align} \omega_{c,\text{TE}} = \frac{c}{2\pi}\sqrt{\frac{(m\pi/a)^2+(n\pi/b)^2}{\mu\epsilon}}, \end{align} \begin{align} \omega_{c,\text{TM}} = \frac{c}{2\pi}\sqrt{\frac{(m\pi/a)^2+(n\pi/b)^2}{\mu\epsilon}}, \end{align} where \(c\) is the speed of light, \(\mu\) and \(\epsilon\) are the permeability and permittivity of the medium, respectively, and \(m\) and \(n\) are the mode indices.
02

Compute the number of modes that have a cutoff frequency less than the given frequency

We are given the frequency \(\omega_{\max}x\). To find the total number of modes that have a cutoff frequency less than this given frequency, we need to find the mode indices \((m, n)\) for both TE and TM modes such that their cutoff frequencies are less than the given frequency. For TE and TM modes, the expression for the cutoff frequencies is the same: $$\omega_c = \frac{c}{2\pi}\sqrt{\frac{(m\pi/a)^2+(n\pi/b)^2}{\mu\epsilon}} \leq \omega_{\max}x.$$ We can solve this inequality to find the values of \(m\) and \(n\). First, let's rearrange the inequality: $$\frac{(m\pi/a)^2+(n\pi/b)^2}{\mu\epsilon} \leq \frac{4\pi^2\omega_{\max}^2 x^2}{c^{-2}}.$$ As we are looking for the number of modes \(N\), we can sum over the mode indices: $$N = \sum_{m=0}^{\infty} \sum_{n=0}^{\infty} (m, n) = \frac{\omega_{\max}^2}{2\pi c^2}ab,$$
03

Compute the density of modes per unit frequency interval

Next, we need to find the derivative of \(N\) with respect to \(\omega\) to compute the density of modes per unit frequency interval, \(dN/d\omega\). Taking the derivative of \(N\) with respect to \(\omega\), we get: $$\frac{dN}{d\omega} = \frac{d}{d\omega} \left( \frac{\omega_{\max}^2}{2\pi c^2} ab \right) = \frac{2\omega_{\max}}{2\pi c^2} ab,$$ where we can see that the density of modes per unit frequency interval increases linearly with frequency, as required.

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Most popular questions from this chapter

Show that the reflection coefficients for the magnetic field amplitudes (either B or H) are identical with \((8.6 .28)\) and \((8.6 .36)\), while the transmission coefficients differ from (8.6.29) and \((8.6 .37)\) by the ratio of the wave impedances of the two media, \((8.5 .18)\) or \((8.5 .19)\). Specifically, show that for the B field, $$ \frac{T_{B}}{T_{\boldsymbol{B}}}=\frac{c_{1}}{c_{2}}=\left(\frac{\kappa_{n k} k_{m 1}}{\kappa_{A 1} k_{m 1}}\right)^{1 / 2}, $$ which is the relative refractive index for the two media; for the \(\mathbf{H}\) field, $$ \frac{T_{H}}{T_{E}}=\frac{Z_{61}}{Z_{42}}=\left(\frac{\kappa_{A 1 \pi_{m 1}}}{\kappa_{A 1 K_{m 2}}}\right)^{1 / 2} $$ Justify the cosine ratio in (8.6.39).

The text following (8.2.10) refers to low-frequency (or dc) laboratory measurements of \(\epsilon_{0}\) and \(\mu_{0}\). How could you determine these constants? What logical chain of definitions and calibrations would be needed?

Consider total reflection at an interface between two nonmagnetic media, with relative refractive index \(n=c_{1} / c_{2}<1\). For angles of incidence \(\theta_{1}\) exceeding the critical angle of (8.6.42), Snell's law gives $$ \sin \theta_{2}=\frac{\sin \theta_{1}}{n}>1, $$ which implies that \(\theta_{2}\) is a complex angle with an imaginary cosine, $$ \cos \theta_{2}=\left(1-\sin ^{2} \theta_{2}\right)^{1 / 2}=j\left(\frac{\sin ^{2} \theta_{1}}{n^{2}}-1\right)^{1 / 2} $$ Substitute these relations in the case I reflection coefficient (8.6.28) to establish $$ R_{\mathbf{E} \perp}=e^{-i 2 \phi_{\perp}}, $$ where $$ \tan \phi_{\perp}=\frac{\left(\sin ^{2} \theta_{1}-n^{2}\right)^{1 / x}}{\cos \theta_{1}} $$ That is, the magnitude of the reflection coefficient is unity, but the phase of the reflected wave depends upon angle. Similarly show for case II from (8.6.36), that \(R_{\text {III }}=e^{-\text {jod with }}\) $$ \tan \phi \|=\frac{\left(\sin ^{2} \theta_{1}-n^{2}\right)^{1 / 2}}{n^{2} \cos \theta_{1}}=\frac{1}{n^{2}} \tan \phi_{\perp} . $$ Note that the two phase shifts are different, so that in general the state of polarization of an incident wave is altered.

It is often convenient to discuss electromagnetic problems in terms of potentials rather than fields. For instance, elementary treatments show that the electrostatic field \(\mathbf{E}(\mathbf{r})\) is conservative and can be derived from a scalar potential function \(\phi(\mathbf{r})\), which is related to \(\mathbf{E}\) by $$ \begin{aligned} &\phi=-\int_{r_{0}}^{r} \mathbf{E} \cdot d \mathbf{l} \\ &\mathbf{E}=-\nabla \phi \end{aligned} $$ Mathematically, the conservative nature of the static field \(\mathbf{E}\) is expressed by the vanishing of its curl. Since the curl of any gradient is identically zero, use of the scalar potential automatically satisfies the static limit of the Maxwell equation (8.2.2); the other constraint on \(\phi\) is Gauss' law (8.2.1). Which hecomes Poisson's equation $$ \nabla^{2} \phi=-\frac{\rho}{\epsilon_{0}} $$ (a) Show that \((8.2 .3)\) is satisfied automatically if we introduce the magnetic vector potential \(\mathbf{A}\), related to the magnetic field by $$ B=\nabla \times A . $$ (b) Show that in the general (nonstatic) case, the electric field is given in terms of the scalar and vector potentials by $$ \mathbf{E}=-\nabla \phi-\frac{\partial \mathbf{A}}{\partial t} $$ (c) Complete the prescription of \(\mathbf{A}\) by defining its divergence by the Lorents condition $$ \boldsymbol{\nabla} \cdot \mathbf{A}=-\frac{1}{c^{2}} \frac{\partial \phi}{\partial t} $$ and show that the two potentials obey the symmetrical inhomogeneous wave equations $$ \begin{aligned} &\nabla^{2} \phi-\frac{1}{c^{2}} \frac{\partial^{2} \phi}{\partial t^{2}}=-\frac{\rho}{\epsilon_{0}} \\ &\nabla^{2} \mathbf{A}-\frac{1}{c^{2}} \frac{\partial^{2} \mathbf{A}}{\partial t^{2}}=-\mu_{0} \mathbf{J} . \end{aligned} $$ These equations connect the potentials associated with radiation fields with their sources \(\rho\) and \(\mathbf{J}\).

Practical coaxial lines used for the distribution of high-frequency signals often consist of a thin copper wire in a polyethylene sleeve on which a copper braid is woven (usually there is also a protective plastic jacket over the braid). Commercial lines are made with nominal characteristic impedances of 50,75 , or 90 ohms. A common 50 -ohm variety has a center conductor of diameter 0035 in. The dielectric constant of polyethylene is \(2.3\) What is the nominal (inside) diameter of the copper braid? What are the capacitance and inductance per foot? What is the speed of propagation, expressed as a percent of the velocity of light? Arswer: \(0120 \mathrm{in} ; 30 \mathrm{pF} / \mathrm{ft} ; 0074 \mu \mathrm{H} / \mathrm{ft} ; 66\) percent.

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